MP2307DN

Part: MP2307 from Monolithic Power Systems (MPS)

Type: Synchronous Buck Converter

Description: A monolithic synchronous buck regulator providing 3A of continuous load current over a 4.75V to 23V input voltage range, with integrated 100mΩ MOSFETs and a fixed 340KHz switching frequency.

Operating Conditions:

• Supply voltage: 4.75V to 23V
• Output voltage range: 0.925V to 20V
• Switching frequency: 340 KHz (fixed)

Absolute Maximum Ratings:

Key Specs:

• Feedback Voltage (VFB): 0.900 V min, 0.925 V typ, 0.950 V max (4.75V ≤ VIN ≤ 23V)
• High-Side Switch On-Resistance (RDS(ON)1): 100 mΩ typ
• Low-Side Switch On-Resistance (RDS(ON)2): 100 mΩ typ
• Upper Switch Current Limit: 4.0 A min, 5.8 A typ
• Oscillation Frequency (Fosc1): 300 KHz min, 340 KHz typ, 380 KHz max
• Shutdown Supply Current: 0.3 μA typ, 3.0 μA max (VEN = 0V)
• Input Under Voltage Lockout Threshold: 3.80 V min, 4.05 V typ, 4.40 V max (VIN Rising)
• Thermal Shutdown: 160 °C typ

Features:

• 3A Continuous Output Current 4A Peak Output Current
• Wide 4.75V to 23V Operating Input Range
• Integrated 100mΩ Power MOSFET Switches
• Output Adjustable from 0.925V to 20V
• Up to 95% Efficiency
• Programmable Soft-Start
• Stable with Low ESR Ceramic Output Capacitors
• Fixed 340KHz Frequency
• Cycle-by-Cycle Over Current Protection
• Input Under Voltage Lockout
• Thermally Enhanced 8-Pin SOIC Package

Applications:

• Distributed Power Systems
• Networking Systems
• FPGA, DSP, ASIC Power Supplies
• Green Electronics/Appliances
• Notebook Computers

Package:

• 8-pin SOIC package

• 3A Continuous Output Current 4A Peak Output Current
• Wide 4.75V to 23V Operating Input Range
• Integrated 100mΩ Power MOSFET Switches
• Output Adjustable from 0.925V to 20V
• Up to 95% Efficiency
• Programmable Soft-Start
• Stable with Low ESR Ceramic Output Capacitors
• Fixed 340KHz Frequency
• Cycle-by-Cycle Over Current Protection
• Input Under Voltage Lockout
• Thermally Enhanced 8-Pin SOIC Package

• Distributed Power Systems
• Networking Systems
• FPGA, DSP, ASIC Power Supplies
• Green Electronics/Appliances
• Notebook Computers

"MPS" and "The Future of Analog IC Technology" are Registered Trademarks of Monolithic Power Systems, Inc.

OPERATION

FUNCTIONAL DESCRIPTION

The MP2307 regulates input voltages from 4.75V to 23V down to an output voltage as low as 0.925V, and supplies up to 3A of load current.

The MP2307 uses current-mode control to regulate the output voltage. The output voltage is measured at FB through a resistive voltage divider and amplified through the internal transconductance error amplifier. The voltage at the COMP pin is compared to the switch current (measured internally) to control the output voltage.

The converter uses internal N-Channel MOSFET switches to step-down the input voltage to the regulated output voltage. Since the high side MOSFET requires a gate voltage greater than the input voltage, a boost capacitor connected between SW and BS is needed to drive the high side gate. The boost capacitor is charged from the internal 5V rail when SW is low.

When the FB pin voltage exceeds 20% of the nominal regulation value of 0.925V, the over voltage comparator is tripped and the COMP pin and the SS pin are discharged to GND, forcing the high-side switch off.

Figure 1—Functional Block Diagram

APPLICATIONS INFORMATION COMPONENT SELECTION

Setting the Output Voltage

The output voltage is set using a resistive voltage divider connected from the output voltage to FB. The voltage divider divides the output voltage down to the feedback voltage by the ratio:

$V_FB = V_OUT frac{R2}{R1 + R2}$

Thus the output voltage is:

$V_OUT = 0.925 × frac{R1 + R2}{R2}$

R2 can be as high as $100k\Omega$ , but a typical value is $10k\Omega$ . Using the typical value for R2, R1 is determined by:

$R1 = 10.81 × (V_OUT - 0.925) (kΩ)$

For example, for a 3.3V output voltage, R2 is $10k\Omega$ , and R1 is $26.1k\Omega$ . Table 1 lists recommended resistance values of R1 and R2 for standard output voltages.

Table 1—Recommended Resistance Values

Inductor

The inductor is required to supply constant current to the load while being driven by the switched input voltage. A larger value inductor will result in less ripple current that will in turn result in lower output ripple voltage. However, the larger value inductor will have a larger physical size, higher series resistance, and/or lower saturation current. A good rule for determining inductance is to allow the peak-topeak ripple current to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit.

The inductance value can be calculated by:

$L = frac{V_OUT}{f_S × Δ I_L} × ≤ft(1 - frac{V_OUT}{V_IN}right)$

Where $V_{OUT}$ is the output voltage, $V_{IN}$ is the input voltage, f_S is the switching frequency, and $\Delta I_1$ is the peak-to-peak inductor ripple current.

Choose an inductor that will not saturate under the maximum inductor peak current, calculated by:

$I_LP = I_LOAD + frac{V_OUT}{2 × f_S × L} × ≤ft(1 - frac{V_OUT}{V_IN}right)$

Where $I_{LOAD}$ is the load current.

The choice of which style inductor to use mainly depends on the price vs. size requirements and any EMI constraints.

Optional Schottky Diode

During the transition between the high-side switch and low-side switch, the body diode of the low-side power MOSFET conducts the inductor current. The forward voltage of this body diode is high. An optional Schottky diode may be paralleled between the SW pin and GND pin to improve overall efficiency. Table 2 lists example Schottky diodes and their Manufacturers.

Table 2—Diode Selection Guide

Input Capacitor

The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors will also suffice. Choose X5R or X7R dielectrics when using ceramic capacitors.

Since the input capacitor (C1) absorbs the input switching current, it requires an adequate ripple current rating. The RMS current in the input capacitor can be estimated by:

$I_C1 = I_LOAD × √{frac{V_OUT}{V_IN}} × ≤ft(1 - frac{V_OUT}{V_IN}right)$

The worst-case condition occurs at $V_{IN} = 2V_{OUT}$ , where $I_{C1} = I_{LOAD}/2$ . For simplification, use an input capacitor with a RMS current rating greater than half of the maximum load current.

The input capacitor can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor, i.e. $0.1\mu F$ , should be placed as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at input. The input voltage ripple for low ESR capacitors can be estimated by:

$Δ V_IN = frac{I_LOAD}{C1 × f_S} × frac{V_OUT}{V_IN} × ≤ft(1 - frac{V_OUT}{V_IN}right)$

Where C1 is the input capacitance value.

Output Capacitor

The output capacitor (C2) is required to maintain the DC output voltage. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. Low ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by:

$Δ V_OUT = frac{V_OUT}{f_S × L} × ≤ft(1 - frac{V_OUT}{V_IN}right) × ≤ft(R_ESR + frac{1}{8 × f_S × C2}right)$

Where C2 is the output capacitance value and $R_{\text{ESR}}$ is the equivalent series resistance (ESR) value of the output capacitor.

When using ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance which is the main cause for the output voltage ripple. For simplification, the output voltage ripple can be estimated by:

$Δ V_OUT = frac{V_OUT}{8 × f_s² × L × C2} × ≤ft(1 - frac{V_OUT}{V_IN}right)$

When using tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to:

$Δ V_OUT = frac{V_OUT}{f_s × L} × ≤ft(1 - frac{V_OUT}{V_IN}right) × R_ESR$

The characteristics of the output capacitor also affect the stability of the regulation system. The MP2307 can be optimized for a wide range of capacitance and ESR values.

Compensation Components

MP2307 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to govern the characteristics of the control system.

The DC gain of the voltage feedback loop is given by:

$A_VDC = R_LOAD × G_CS × A_EA × frac{V_FB}{V_OUT}$

Where $V_{FB}$ is the feedback voltage (0.925V), $A_{VEA}$ is the error amplifier voltage gain, $G_{CS}$ is the current sense transconductance and $R_{LOAD}$ is the load resistor value.

The system has two poles of importance. One is due to the compensation capacitor (C3) and the output resistor of the error amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at:

$f_P1 = frac{G_EA}{2π × C3 × A_VEA}$

$f_P2 = frac{1}{2π × C2 × R_LOAD}$

Where G_EA is the error amplifier transconductance.

The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:

$f_Z1 = frac{1}{2π × C3 × R3}$

The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero, due to the ESR and capacitance of the output capacitor, is located at:

$f_ESR = frac{1}{2π × C2 × R_ESR}$

In this case, a third pole set by the (C6) compensation capacitor and compensation resistor (R3) is used compensate the effect of the ESR zero on the loop gain. This pole is located at:

$f_P3 = frac{1}{2π × C6 × R3}$

The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is important. Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies could cause system instability. A good standard is to set the crossover frequency below one-tenth of the switching frequency.

To optimize the compensation components, the following procedure can be used.

1. Choose the compensation resistor (R3) to set the desired crossover frequency.

Determine R3 by the following equation:

$R3 = frac{2π × C2 × f_C}{G_EA × G_CS} × frac{V_OUT}{V_FB} < frac{2π × C2 × 0.1 × f_S}{G_EA × G_CS} × frac{V_OUT}{V_FB}$

Where f_C is the desired crossover frequency which is typically below one tenth of the switching frequency.

1. Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero $(f_{Z1})$ below one-forth of the crossover frequency provides sufficient phase margin.

Determine C3 by the following equation:

$C3 > frac{4}{2π × R3 × f_C}$

Where R3 is the compensation resistor.

1. Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the switching frequency, or the following relationship is valid:

$frac{1}{2π × C2 × R_ESR} < frac{f_S}{2}$

If this is the case, then add the second compensation capacitor (C6) to set the pole f_P3 at the location of the ESR zero. Determine C6 by the equation:

$C6 = frac{C2 × R_ESR}{R3}#### PCB Layout Guide

PCB layout is very important to achieve stable operation. It is highly recommended to duplicate EVB layout for optimum performance.

If change is necessary, please follow these guidelines and take Figure for reference.

• Keep the path of switching current short and minimize the loop area formed by Input cap., high-side MOSFET and low-side MOSFET.
• Bypass ceramic capacitors are suggested to be put close to the Vin Pin.
• 1. Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close to the chip as possible.
• 1. R_OUT SW away from sensitive analog areas such as FB.

1. Connect IN, SW, and especially GND respectively to a large copper area to cool the chip to improve thermal performance and long-term reliability.

TOP Layer

Bottom Layer

Figure 2—PCB Layout (Double Layer)

External Bootstrap Diode

An external bootstrap diode may enhance the efficiency of the regulator, the applicable conditions of external BS diode are:

• V_OUT is 5V or 3.3V; and
• Duty cycle is high:D = \frac{V_{OUT}}{V_{IN}} > 65%In these cases, an external BS diode is recommended from the output of the voltage regulator to BS pin, as shown in Figure 3

Figure 3—Add Optional External Bootstrap Diode to Enhance Efficiency

The recommended external BS diode is IN4148, and the BS cap is0.1\sim1\mu$ F.

VIN = 12V, TA = +25°C, unless otherwise noted.